Wireless communications may suffer from four major impairments, namely path loss, multipath fading, inter-symbol interference and co-channel interference. Adaptive antenna systems may be used to suppress the effects of such impairments to improve the performance of wireless communication systems. There are two types of adaptive antenna systems, namely diversity antenna systems and beamforming antenna systems. In a diversity antenna system, multiple low-correlation or independent fading channels are utilised in order to compensate multipath fading, thus achieving diversity gain. Beamforming antenna systems, on the other hand, achieve beamforming gain by applying spatial directivity, thus limitedly compensating for path loss and suppressing co-channel interference.
In a diversity antenna system, antenna spacing is usually required to be sufficiently large, for example 10λ, in order to achieve low-correlation or independent fading channels. This is especially true for small angular spread environments. However, a beamforming antenna system need to achieve spatial directivity, so signals received at and/or transmitted from all beamforming antennas must be correlated. This means that for a beamforming antenna system, antenna spacing is usually small, for example half wavelength for a uniform linear array (ULA). Because of the conflict between the required antenna spacing for diversity and beamforming antenna systems, there exists a technical prejudice among those skilled in the art that diversity and beamforming gains cannot be achieved simultaneously.
A classical approach to achieving antenna diversity is to use multiple antennas at the receiver and perform combining or selection to improve the quality of the received signal.
The major problems faced with using the receiving antenna diversity or diversity reception approach in the downlink of wireless communication systems relate to constraints in cost, size and power consumption of the receiver. For apparent reasons, features such as small size, lightweight and low cost are paramount considerations in designing wireless communications receivers. The addition of multiple antennas and radio frequency (RF) chains or selection and switching circuits in such receivers is presently not feasible. Consequently, diversity reception techniques are often applied only to improve the uplink transmission quality by the use of multiple antennas at the base station. Since a base station often serves thousands of receivers, it is more economical to add equipment to the base station rather than the receivers.
Other approaches relating to transmitting antenna diversity or diversity transmission have also been suggested, for instance, a delay diversity scheme is proposed by A. Wittneben in “A new bandwidth efficient transmitting antenna modulation diversity scheme for linear digital modulation”, Proceeding of the 1993 IEEE International Conference on Communications (ICC'93), pp. 1630-1634, May 1993. This proposal relates to a base station transmitting a sequence of symbols through one antenna and the same but delayed sequence of symbols through another antenna. A special case of this scheme was proposed by N. Seshadri and J. H. Winters in “Two signaling schemes for improving the error performance of frequency-division-duplex (FDD) transmission systems using transmitting antenna diversity”, International Journal of Wireless Information Networks, Vol. 1, No. 1, 1994. The sequence of codes in this scheme is routed through a cycling switch that directs each code to various antennas in succession. Since copies of the same symbols are transmitted through multiple antennas at different times, both space and time diversities are achieved.
A coding technique known as space-time trellis coding (STTC) has been proposed by V.Tarokh, N. Seshadri and A. R. Calderbank in “Space-time codes for high data rate wireless communication: Performance analysis and code construction”, IEEE trans. on Information Theory, vol. 44, No. 3, pp. 744-765, March 1998. The STTC technique combines signal processing at the receiver with coding techniques appropriate to multiple transmitting antennas, and provides significant gain over aforementioned approaches. Specific space-time trellis codes designed for two to four transmitting antennas perform well in slow fading environments, which is typical of indoor transmission, and come within 2-3 dB of the theoretical outage capacity. The bandwidth efficiency achieved is about three to four times of other systems. The space-time trellis codes proposed provide the best possible trade-off between constellation size, data rate, diversity advantage, and trellis complexity. However, when the number of transmitting antennas is fixed, the decoding complexity of the STTC technique, as measured by the number of trellis states at the decoder, increases exponentially as a function of both diversity level and transmission rate.
In attempting to further improve downlink performance where the STTC technique is employed, a scheme to combine a beamforming technique with space-time trellis coding is proposed by R. Negi, A. M. Tehrani and J. Cioffi in “Adaptive antennas for space-time coding over block invariant multipath fading channels”, Proc. of IEEE VTC, pp. 70-74, 1999. In the proposal, an optimum beamformer is derived based on the coding criteria given in the proposal by V. Tarokh et al to achieve diversity gain as well as beamforming gain. This scheme achieves diversity gain and beamforming gain simultaneously for wireless communication systems employing the STTC technique. For a given achievable diversity gain, optimal transmission beamforming weights to maximize beamforming gain are found via singular-value decomposition (SVD) based on coding techniques.
For addressing the issue of decoding complexity of the STTC technique, S. M. Alamouti proposes a scheme for transmission using two antennas in “A simple diversity transmission technique for wireless communications”, IEEE Journal of Selected Areas in Communications, Vol. 16, No.8, pp.1451-1458, October 1998. This involves a maximum likelihood detection scheme based only on linear processing at the receiver. A space-time block coding (STBC) technique introduced by V. Tarokh, N. Seshadri, and A. R. Calderbank in “Space-time block codes from orthogonal designs,” IEEE Trans. On Information Theory, Vol. 45, pp. 1456-1467, July 1999, generalizes this transmission scheme to an arbitrary number of transmitting antennas and is able to achieve the full diversity promised by the transmitting and receiving antennas. Unfortunately, higher order complex orthogonal block codes all have less than unity coding rate, which results in a reduction in data throughput or an expansion in bandwidth in order to maintain the same date rate. Furthermore, it is not clear whether using higher order diversity transmission directly or applying other error correction codes on top of the second order diversity transmission system achieves better overall performance. Therefore, the two-antenna STBC scheme proposed by Alamouti remains one of the most attractive schemes for its simplicity and unity coding-rate.
FIG. 1 is a block diagram of a wireless communication system for illustrating Alamouti's proposed diversity scheme using two transmitting antennas 102 equipped at a base station (BS) 104 for achieving diversity transmission. A signal s(n) to be transmitted is first coded in a space-time block coding module 106 in the base station 104 which has one input port 108 and two output ports 110. The input port 108 accepts the transmitted sequence, s(0), s(1), . . . s(n) and the two output ports 110 provide, in response, respective output signals s1(t) and s2(t) at time instants t=n and t=n+1 as follows:
t = nt = n + 1s1(t)s(n)/{square root over (2)}s* (n + 1)/{square root over (2)}s2(t)s(n + 1)/{square root over (2)}−s* (n)/{square root over (2)}where n is an even integer and ‘*’ denotes a complex conjugate operation.
At a single receiving antenna 112 at a mobile terminal 114, signals received at time instants t=n and t=n+1 are given by:x(n)=α1s1(n)+α2s2(n)+v(n)  (1)x(n+1)=α1s1(n+1)+α2s2(n+1)+v(n+1)  (2)where α1 and α2 are the respective channel coefficients 116 from the two transmitting antennas 102 to the receiving antenna 112, respectively, and v(n) is the additive white Gaussian noise (AWGN).
The received signal is subsequently decoded by a space-time decoding module 118 in the mobile terminal 114. Specifically, equations (1) and (2) may be written in matrix forms:                               [                                                                      x                  ⁡                                      (                    n                    )                                                                                                                        x                  ⁡                                      (                                          n                      +                      1                                        )                                                                                ]                =                                                            1                                  2                                            ⁡                              [                                                                                                    s                        ⁡                                                  (                          n                          )                                                                                                                                    s                        ⁡                                                  (                                                      n                            +                            1                                                    )                                                                                                                                                                                                  s                          *                                                ⁡                                                  (                                                      n                            +                            1                                                    )                                                                                                                                    -                                                                              s                            *                                                    ⁡                                                      (                            n                            )                                                                                                                                              ]                                      ⁡                          [                                                                                          α                      1                                                                                                                                  α                      2                                                                                  ]                                +                      [                                                                                v                    ⁡                                          (                      n                      )                                                                                                                                        v                    ⁡                                          (                                              n                        +                        1                                            )                                                                                            ]                                              (        3        )                                          [                                                                      x                  ⁡                                      (                    n                    )                                                                                                                                            x                    *                                    ⁡                                      (                                          n                      +                      1                                        )                                                                                ]                =                                                            1                                  2                                            ⁡                              [                                                                                                    α                        1                                                                                                            α                        2                                                                                                                                                -                                                  α                          2                          *                                                                                                                                    α                        1                        *                                                                                            ]                                      ⁡                          [                                                                                          s                      ⁡                                              (                        n                        )                                                                                                                                                        s                      ⁡                                              (                                                  n                          +                          1                                                )                                                                                                        ]                                +                      [                                                                                v                    ⁡                                          (                      n                      )                                                                                                                                                              v                      *                                        ⁡                                          (                                              n                        +                        1                                            )                                                                                            ]                                              (        4        )            
Therefore, channel coefficients 116 may be estimated via equation (3) using training symbols, while equation (4) may be used for signal estimation/detection. In such a scheme proposed by Alamouti, channel coefficient estimation and signal detection involve very simple linear operations. Also, compared to a diversity reception technique involving one transmitting antenna and two receiving antennas, the scheme achieves the same order of diversity gain as diversity reception techniques using a maximum ratio combining (MRC) approach, although the scheme suffers from a 3 dB performance loss.
To achieve diversity gain and beamforming gain simultaneously for systems employing the STBC technique, it is apparent that the transmission structure adopted by STTC systems to achieve the both gains can be employed similarly. But it is not very clear for STBC systems how much these two kinds of gain can be obtained and how to determine beamforming weights. Therefore a method to obtain the optimal beamforming weights, in terms of obtaining fall diversity as well as maximizing output signal-to-noise ratio, is derived in hereinafter from a signal processing point of view.
FIG. 2 shows a system employing the STBC technique, which combines beamforming technique with diversity transmission. A signal to be transmitted, s(n), is first provided to an input 202 of a space-time block encoder 204 and coded by it yielding two branch outputs 206 as s1(n) and s2(n). The output signals are then passed into two transmit beamformers w1 208 and w2 210 for beamforming processing, and passed on to a signal combiner 212 which performs a simple summing function of the two beamforming processed inputs to produce a vector signal x(n) 214 for transmission through multiple antennas 216. The signal x(n) 214 may be expressed as:                               x          ⁡                      (            n            )                          =                                            w              1              H                        ⁢                                          s                1                            ⁡                              (                n                )                                              +                                    w              2              H                        ⁢                                          s                2                            ⁡                              (                n                )                                                                        (        5        )            
The physical channel is assumed to consist of L spatially separated paths, of which fading coefficients and directions of arrival (DOAs) are denoted as (αl(t),θl) for l=1,Λ,L. If the maximum time delay relative to the first arrived path is smaller than the symbol interval, a flat fading channel is observed, and the instantaneous channel response 218 may be expressed as:                               h          ⁡                      (            t            )                          =                              ∑                          l              =              1                        L                    ⁢                                           ⁢                                                    α                l                            ⁡                              (                t                )                                      ·                          a              ⁡                              (                                  θ                  l                                )                                                                        (        6        )            where a(θl) is the downlink steering vector at DOA θl. The signal y(n) arriving at a receiving antenna 220 at a mobile terminal 222 is given by:                               y          ⁡                      (            n            )                          =                                            w              1              H                        ·                          h              ⁡                              (                t                )                                      ·                                          s                1                            ⁡                              (                n                )                                              +                                    w              2              H                        ·                          h              ⁡                              (                t                )                                      ·                                          s                2                            ⁡                              (                n                )                                              +                      v            ⁡                          (              n              )                                                          (        7        )            
By denoting                     β        1            ⁡              (        t        )              =                                        w            1            H                    ·                      h            ⁡                          (              t              )                                      ⁢                                   ⁢        and        ⁢                                   ⁢                              β            2                    ⁡                      (            t            )                              =                        w          2          H                ·                  h          ⁡                      (            t            )                                ,transmission beamforming weights may be estimated by maximizing the cost function:J=E[|β1(t)|2+|β2(t)|2]  (8)s.t. E[β1(t)·β2*(t)]=0  (9)and                                                         w              1              H                        ·                          w              1                                +                                    w              2              H                        ·                          w              2                                      =        1                            (        10        )            where ‘E’ denotes the expectation operation. Maximum SNR is obtained by maximizing (8) subject to (10), while condition (9) guarantees that β1(t) and β2(t) are statistically uncorrelated thus achieving full diversity gain.
Comparing (1) with (7), with the aid of downlink beamforming, two statistical uncorrelated fading channels, β1(t) and β2 (t) have been artificially generated, with which space-time block decoding may be used to recover the transmitted signal s(n).
The optimal transmission beamforming weight vectors are Eigenvectors corresponding to the two largest Eigenvalues of the downlink channel covariance matrix:
 R=E[h(t)·hH(t)]  (11)
where the expectation is conducted over all fading coefficients. Suppose the average power of each path is E|αl(t)|2=γl, the covariance matrix is given by:                     R        =                              ∑                          l              =              1                        L                    ⁢                                           ⁢                                    γ              l                        ·                          a              ⁡                              (                                  θ                  l                                )                                      ·                                          a                H                            ⁡                              (                                  θ                  l                                )                                                                        (        12        )            
To achieve full diversity and maximized beamforming, this scheme requires only up to second order statistics of fading channel. Whereas the scheme involving the combining of the STTC technique with beamforming, beamforming weights are guaranteed to be optimal for Rayleigh/Rice fading channel only.
The aforementioned schemes for systems employing either the STTC or STBC technique achieve diversity gain and optimal beamforming gain via combining the space-time coding techniques with the adaptive beamforming technique. The optimal beamforming weights may be obtained through either SVD or Eigen-decomposition. However, theoretical performance benefits achieved by the adaptive systems may be offset by the cost and complexities encountered in implementation. In wireless communications environments, received complex envelopes of signals vary rapidly due to fading. The adaptive beamformer must therefore “track” the change in the fading envelope so that beamforming weights always meet the optimality requirements. If the tracking performance using the adaptive algorithm is not sufficient, beamforming performances may degrade.
For the beamforming technique, one simple alternative to the adaptive antenna system is the switched beam antenna system. The switched beam antenna system consists of multiple narrow beams, of which the beam considered best or has the strongest received power is used to serve a desired mobile terminal as it is moved through the coverage of a base station. The switched beam antenna system is more economical to implement, requiring only a static beamforming network, RF switch and control. Commercial deployments of switched beam antenna systems have demonstrated significant performance improvements over systems without multiple antennas. Nevertheless, the performance of the switched beam antenna system is considered by those skilled in the art to be inferior to that of the adaptive antenna system. Hence, because of the trade-off between implementation complexity and system performance, there is a technical prejudice that directly combining a switched beam antenna system with a space-time coding technique sacrifices system performance for the complexity of implementation.
Therefore, there is a need for a system that provides the performance of an adaptive antenna system while possessing the implementation simplicity of a switched beam system for achieving diversity gain and beamforming gain simultaneously.